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| L6711 3 PHASE CONTROLLER WITH DYNAMIC VID AND SELECTABLE DACs 1 FEATURES 2A INTEGRATED GATE DRIVERS FULL DIFFERENTIAL CURRENT READING ACROSS INDUCTOR OR LOW-SIDE MOSFET 0.5% OUTPUT VOLTAGE ACCURACY 6 BIT PROGRAMMABLE OUTPUT FROM 0.8185V TO 1.5810V IN 12.5mV STEPS 5 BIT PROGRAMMABLE OUTPUT FROM 0.800V TO 1.550V IN 25mV STEPS DYNAMIC VID MANAGEMENT ADJUSTABLE REFERENCE VOLTAGE OFFSET 3% ACTIVE CURRENT SHARING ACCURACY DIGITAL 2048 STEP SOFT-START PROGRAMMABLE OVERVOLTAGE PROTECTION INTEGRATED TEMPERATURE SENSOR CONSTANT OVER CURRENT PROTECTION OSCILLATOR INTERNALLY FIXED AT 150kHz (450kHz RIPPLE) OSCILLATOR EXTERNALLY ADJUSTABLE OUTPUT ENABLE INTEGRATED REMOTE SENSE BUFFER TQFP48 7x7 PACKAGE WITH EXPOSED PAD Figure 1. Package TQFP48 (exposed pad) Table 1. Order Codes Part Number L6711 L6711TR Package TQFP48 TQFP48 in Tape & Reel 3 DESCRIPTION The device implements a three phase step-down controller with a 120 phase-shift between each phase with integrated high current drivers in a compact 7x7mm body package with exposed pad. The device embeds selectable DAC: the output voltage ranges from 0.8185V to 1.5810V with 12.5mV steps (VID_SEL=OPEN) or from 0.800V to 1.550V with 25mV steps (VID_SEL=GND; VID5 drives an optional +25mV offset) managing dynamic VID with 0.5% accuracy over line and temp variations. Additional programmable offset can be added to the voltage reference with a single external resistor. The device assures a fast protection against load over current and load over/under voltage. An internal crowbar is provided turning on the low side mosfet if an over-voltage is detected. In case of over-current, the system works in Constant Current mode until UVP. Selectable current reading adds flexibility in system design. 2 APPLICATIONS HIGH CURRENT VRM/VRD FOR DESKTOP / SERVER / WORKSTATION CPUs HIGH DENSITY DC/DC CONVERTERS November 2004 Rev. 2 1/38 L6711 Figure 2. Block Diagram VCCDR1 VCCDR2 VCCDR3 PHASE1 PHASE2 PHASE3 UGATE1 UGATE2 UGATE3 LGATE1 LGATE2 LGATE3 PGND1 PGND2 PGND3 12.5A BOOT1 BOOT2 BOOT3 HS3 OUTEN OUTEN HS1 HS1 LOGIC PWM ADAPTIVE ANTI CROSS CONDUCTION HS2 LS2 LOGIC PWM ADAPTIVE ANTI CROSS CONDUCTION CURRENT SHARING CORRECTION LS3 LOGIC PWM ADAPTIVE ANTI CROSS CONDUCTION CURRENT SHARING CORRECTION OVP OVP VCC VCC SGND 3 PHASE OSCILLATOR OSC CURRENT SHARING CORRECTION PWM1 PWM2 PWM3 CS_SEL TC PWM1 DIGITAL SOFT START VCC TEMPERATURE COMPENSATION OUTEN PWM2 PWM3 AVERAGE CURRENT CH3 CURRENT READING OCP3 CH2 CURRENT READING OCP2 CS3CS3+ VCCDR CS_SEL L6711 CONTROL LOGIC AND PROTECTIONS OCP1 OCP2 OCP3 CS2CS2+ VID0 VID1 VID2 VID3 VID4 VID5 VID_SEL DAC WITH DYNAMIC VID CONTROL OFFSET TOTAL CURRENT IOS IDROOP ITC 115% / OVP 64k 64k ERROR AMPLIFIER REMOTE BUFFER IFB 64k 64k CH1 CURRENT READING OCP1 CS1CS1+ OVP SS_END 1.240V IOS FB FBR OFFSET Table 2. Absolute Maximum Ratings Symbol VCC, VCCDRx VBOOTx-VPHASEx VUGATEx-VPHASEx LGATEx, PHASEx to PGNDx VID0 to VID5 All other pins to PGNDx VPHASEx CS3- Pin OTHER PINS Sustainable Positive Peak Voltage. T<20nS @ 600kHz Maximum Withstanding Voltage Range Test Condition: CDF-AEC-Q100-002"Human Body Model" Acceptance Criteria: "Normal Performance" To PGNDx Boot Voltage Parameter Value 15 15 15 -0.3 to Vcc+0.3 -0.3 to 5 -0.3 to 7 26 1500 2000 Unit V V V V V V V V V Table 3. Thermal Data Symbol Rth j-amb TMAX Tstg Tj Ptot Parameter Thermal Resistance Junction to Ambient (Device soldered on 2s2p PC Board) Maximum junction temperature Storage temperature range Junction Temperature Range Max power dissipation at Tamb = 25C Value 40 150 -40 to 150 0 to 125 2.5 Unit C / W C C C W 2/38 COMP VSEN FBG L6711 Figure 3. Pin Connections (Top view) VID_SEL SS_END PHASE3 UGATE3 LGATE3 PGND3 BOOT3 SGND VID4 VID3 VID2 N.C. VCCDR3 PGND2 LGATE2 VCCDR2 PHASE2 UGATE2 BOOT2 BOOT1 UGATE1 PHASE1 N.C. 36 35 34 33 32 31 30 29 28 27 26 25 37 24 38 39 40 41 42 43 44 45 46 47 48 1 VCCDR1 2 LGATE1 3 PGND1 4 VCC 5 SGND 6 OFFSET 7 COMP 8 FB 9 10 11 12 FBR OVP VSEN FBG 23 22 21 20 VID1 VID0 VID5 OSC / FAULT CS_SEL TC CS3+ CS3CS2+ CS2CS1+ CS1OUTEN L6711 19 18 17 16 15 14 13 Table 4. Pin Function N 1 Name Description VCCDR1 Channel 1 LS driver supply: it can be varied from 5V to 12V buses. It must be connected together with other VCCDRx pins. Filter locally with at least 1F ceramic cap vs. PGND1. LGATE1 PGND1 VCC SGND Channel 1 LS driver output. A little series resistor helps in reducing device-dissipated power. Channel 1 LS driver return path. Connect to Power Ground Plane. Device supply voltage. The operative supply voltage is 12V 15%. Filter with 1F capacitor (Typ.) vs. SGND. All the internal references are referred to this pin. Connect it to the PCB signal ground. 2 3 4 5 6 OFFSET Offset programming pin, internally fixed at 1.240V. Short to SGND to disable the offset generation or connect through a resistor ROFFSET to SGND to program an offset (positive or negative, depending on TC status) to the regulated output voltage as reported in the relative section. COMP FB This pin is connected to the error amplifier output and is used to compensate the control feedback loop. This pin is connected to the error amplifier inverting input and is used to compensate the voltage control feedback loop. Connecting a resistor between this pin and VSEN pin allows programming the droop effect. Over Voltage protection setup pin: it allows programming the OVP intervention. Internally pulled-up to 5V, it sources a constant 12.5A current. Leaving the pin floating the OVP threshold is set to 115% (Typ.) of the programmed voltage Connecting a resistor ROVP to SGND, it sets the OVP threshold to a fixed programmable voltage (see relevant section for further details). Filter with 10nF vs. SGND in this case. Manages Over&Under-voltage conditions. It is internally connected with the output of the Remote Sense Buffer for Remote Sense of the regulated voltage. If no Remote Sense is implemented, connect it directly to the regulated voltage in order to manage OVP and UVP. 7 8 9 OVP 10 VSEN 3/38 L6711 Table 4. Pin Function (continued) N 11 12 13 Name FBR FBG OUTEN Description Remote sense buffer non-inverting input. It has to be connected to the positive side of the load to perform a remote sense. Remote sense buffer inverting input. It has to be connected to the negative side of the load to perform a remote sense. Output Enable pin, internally 3V pulled-up. If forced to a voltage lower than 0.3V, the device stops operation with all mosfets OFF: all the protections are disabled in this condition except pre-OVP. Cycle this pin to recover latch from protections; filter with 1nF (Typ.) capacitor vs. SGND. Channel 1 Current Sense Negative Input pin. It must be connected through an Rg resistor to the LS mosfet drain (or to the LS-side of the sense resistor placed in series to the LS mosfet) if LS mosfet sense is performed (CS_SEL=OPEN). Otherwise (CS_SEL=SGND), it must be connected to the output-side of the output inductor (or the output-side of the sense resistor used and placed between the channel 1 inductor and the output of the converter) through Rg resistor. The net connecting the pin to the sense point must be routed as close as possible to the CS1+ net in order to couple in common mode any picked-up noise. Channel 1 Current Sense Positive Input pin. It must be connected through an Rg resistor to the LS mosfet source (or to the GND-side of the sense resistor placed in series to the LS mosfet) if LS mosfet sense is performed (CS_SEL=OPEN). Otherwise (CS_SEL=SGND), it must be connected to the phase-side of the output inductor (or the inductor-side of the sense resistor used and placed between the channel 1 inductor and the output of the converter) through Rg resistor and an R-C network across the inductor. The net connecting the pin to the sense point must be routed as close as possible to the CS1net in order to couple in common mode any picked-up noise. Channel 2 Current Sense Negative Input pin. It must be connected through an Rg resistor to the LS mosfet drain (or to the LS-side of the sense resistor placed in series to the LS mosfet) if LS mosfet sense is performed (CS_SEL=OPEN). Otherwise (CS_SEL=SGND), it must be connected to the output-side of the output inductor (or the output-side of the sense resistor used and placed between the channel 2 inductor and the output of the converter) through Rg resistor. The net connecting the pin to the sense point must be routed as close as possible to the CS2+ net in order to couple in common mode any picked-up noise. Channel 2 Current Sense Positive Input pin. It must be connected through an Rg resistor to the LS mosfet source (or to the GND-side of the sense resistor placed in series to the LS mosfet) if LS mosfet sense is performed (CS_SEL=OPEN). Otherwise (CS_SEL=SGND), it must be connected to the phase-side of the output inductor (or the inductor-side of the sense resistor used and placed between the channel 2 inductor and the output of the converter) through Rg resistor and an R-C network across the inductor. The net connecting the pin to the sense point must be routed as close as possible to the CS2net in order to couple in common mode any picked-up noise. Channel 3 Current Sense Negative Input pin. It must be connected through an Rg resistor to the LS mosfet drain (or to the LS-side of the sense resistor placed in series to the LS mosfet) if LS mosfet sense is performed (CS_SEL=OPEN). Otherwise (CS_SEL=SGND), it must be connected to the output-side of the output inductor (or the output-side of the sense resistor used and placed between the channel 3 inductor and the output of the converter) through Rg resistor. The net connecting the pin to the sense point must be routed as close as possible to the CS3+ net in order to couple in common mode any picked-up noise. 14 CS1- 15 CS1+ 16 CS2- 17 CS2+ 18 CS3- 4/38 L6711 Table 4. Pin Function (continued) N 19 Name CS3+ Description Channel 3 Current Sense Positive Input pin. It must be connected through an Rg resistor to the LS mosfet source (or to the GND-side of the sense resistor placed in series to the LS mosfet) if LS mosfet sense is performed (CS_SEL=OPEN). Otherwise (CS_SEL=SGND), it must be connected to the phase-side of the output inductor (or the inductor-side of the sense resistor used and placed between the channel 3 inductor and the output of the converter) through Rg resistor and an R-C network across the inductor. The net connecting the pin to the sense point must be routed as close as possible to the CS3net in order to couple in common mode any picked-up noise. Temperature Compensation pin. Connect through a resistor RTC and filter with 10nF vs. SGND to program the temperature compensation effect. Short to SGND to disable the compensation effect. 20 TC 21 CS_SEL Current Reading Selection pin, internally 5V pulled-up. Leave floating to sense current across low-side mosfets or a sense resistor placed in series to the LS mosfet source. Maximum duty cycle is dynamically limited and Track&Hold is enabled to assure proper reading of the current. Short to SGND to read current across inductors or a sense resistor placed in series to the output inductors. No duty cycle limitation and no Track&Hold performed in this case. OSC/ FAULT Oscillator pin. It allows programming the switching frequency of each channel: the equivalent switching frequency at the load side results in being tripled. Internally fixed at 1.24V, the frequency is varied proportionally to the current sunk (forced) from (into) the pin with an internal gain of 6kHz/A (See relevant section for details). If the pin is not connected, the switching frequency is 150kHz for each channel (450kHz on the load). The pin is forced high (5V Typ.) when an Over/Under Voltage is detected; to recover from this condition, cycle VCC or the OUTEN pin. Voltage IDentification pins. Internally pulled-up to 3V, connect to SGND to program a `0' while leave floating to program a `1'. They are used to program the output voltage as specified in Table 5 and Table 6 together with VID_SEL and to set the OVP/UVP protection thresholds accordingly. See relevant section for details about DAC selection. 22 23, 24 to 28 VID5, VID0-4 29 VID_SEL VID_SELect pin. Through this pin it is possible to select the DAC table used for the regulation. Leave floating to use a VRD10.x compliant DAC (See Table 1) while short to SGND to use a VRM-Hammer compliant DAC (See Table 2). See relevant section for details about DAC selection. SS_END Soft start end signal. It is an open collector output, set free after finishing the soft start. Pull-up with a resistor to a voltage lower than 5V, if not used may be left floating. SGND All the internal references are referred to this pin. Connect it to the PCB signal ground. PHASE3 Channel 3 HS driver return path. It must be connected to the HS3 mosfet source and provides the return path for the HS driver of channel 3. UGATE3 Channel 3 HS driver output. A little series resistor helps in reducing device-dissipated power. BOOT3 Channel 3 HS driver supply. This pin supplies the relative high side driver. Connect through a capacitor (100nF Typ.) to the PHASE3 pin and through a diode to VCC (cathode vs. boot). Channel 3 LS driver return path. Connect to Power Ground Plane. Channel 3 LS driver output. A little series resistor helps in reducing device-dissipated power. Not internally connected. 30 31 32 33 34 35 36 37 PGND3 LGATE3 N.C. 5/38 L6711 Table 4. Pin Function (continued) N 38 Name Description VCCDR3 Channel 3 LS driver supply: it can be varied from 5V to 12V buses. It must be connected together with other VCCDRx pins. Filter locally with at least 1F ceramic cap vs. PGND3. PGND2 LGATE2 Channel 2 LS driver return path. Connect to Power Ground Plane. Channel 2 LS driver output. A little series resistor helps in reducing device-dissipated power. 39 40 41 VCCDR2 Channel 2 LS driver supply: it can be varied from 5V to 12V buses. It must be connected together with other VCCDRx pins. Filter locally with at least 1F ceramic cap vs. PGND2. PHASE2 Channel 2 HS driver return path. It must be connected to the HS2 mosfet source and provides the return path for the HS driver of channel 2. UGATE2 Channel 2 HS driver output. A little series resistor helps in reducing device-dissipated power. BOOT2 Channel 2 HS driver supply. This pin supplies the relative high side driver. Connect through a capacitor (100nF Typ.) to the PHASE2 pin and through a diode to VCC (cathode vs. boot). Channel 1 HS driver supply. This pin supplies the relative high side driver. Connect through a capacitor (100nF Typ.) to the PHASE1 pin and through a diode to VCC (cathode vs. boot). 42 43 44 45 BOOT1 46 47 48 PAD UGATE1 Channel 1 HS driver output. A little series resistor helps in reducing device-dissipated power. PHASE1 Channel 1 HS driver return path. It must be connected to the HS1 mosfet source and provides the return path for the HS driver of channel 1. N.C. THERM AL PAD Not internally connected. Thermal pad connects the silicon substrate and makes a good thermal contact with the PCB to dissipate the power necessary to drive the external mosfets. Connect to the GND plane with several vias to improve thermal conductivity. Electrical Characteristcs (VCC=12V15%, TJ = 0C to 70C unless otherwise specified) Symbol VCC SUPPLY CURRENT ICC ICCDRx IBOOTx VCC supply current VCCDRx supply current BOOTx supply current HGATEx and LGATEx open VCCDRx=VBOOTx=12V LGATEx open; VCCDRx=12V HGATEx open;PHASEx to PGNDx VCC=BOOTx=12V 18 1.5 1 23 2 1.5 mA mA mA Parameter Test Condition Min. Typ. Max. Unit POWER-ON Turn-On VCC threshold Turn-Off VCC threshold Turn-On VCCDRx Threshold VCC Rising; VCCDRx=5V VCC Falling; VCCDRx=5V VCCDRx Rising VCC=12V 8.2 6.5 4.2 9.2 7.5 4.4 10.2 8.5 4.6 V V V 6/38 L6711 Electrical Characteristcs (continued) (VCC=12V15%, TJ = 0C to 70C unless otherwise specified) Symbol Parameter Turn-Off VCCDRx Threshold OSCILLATOR AND INHIBIT fOSC OUTENIL OUTENIH dMAX Maximum duty cycle Initial Accuracy Output Enable Threshold OSC = OPEN OSC = OPEN; TJ=0 to 125C Input Low Input High OSC = OPEN: CS_SEL = OPEN; IFB=0 OSC = OPEN; CS_SEL = OPEN; IFB=105A Vosc FAULT Ramp Amplitude Voltage at pin OSC OVP or UVP Active 4.70 0.5 72 30 80 40 3 5.0 5.30 135 127 150 165 178 0.3 kHz kHz V V % % V V Test Condition VCCDRx Falling VCC=12V Min. 4.0 Typ. 4.2 Max. 4.4 Unit V REFERENCE AND DAC Output Voltage Accuracy VRD10.x DAC FBR = VOUT; FBG = GNDOUT HAMMER DAC FBR = VOUT; FBG = GNDOUT IVID, IVID_SEL VID, VID_SEL pull-up Current VID, VID_SEL pull-up Voltage VIDIL VIDIH ERROR AMPLIFIER A0 SR DC Gain Slew-Rate COMP = 10pF 80 15 dB V/s VID, VID_SEL Threshold VIDx = GND VID_SEL=GND VIDx = OPEN VID_SEL = OPEN Input Low Input High 0.8 -0.5 -0.6 3 4.5 3 0.4 0.5 0.6 6 % % A V V V DIFFERENTIAL AMPLIFIER (REMOTE BUFFER) DC Gain CMRR SR Common Mode Rejection Ratio Slew Rate VSEN = 10pF 1 40 15 V/V dB V/s DIFFERENTIAL CURRENT SENSING AND OFFSET ICSxICSx+ Bias Current Bias Current -3 ILOAD = 0 50 50 3 A A *% I INF O x - I A VG Current Sense Mismatch ---------------------------------I AVG IOCTH Over Current Threshold ICSx-(OCP)-ICSx-(0) 30 35 40 A 7/38 L6711 Electrical Characteristcs (continued) (VCC=12V15%, TJ = 0C to 70C unless otherwise specified) Symbol IFB Parameter Droop Current deviation from nominal value Offset Current Test Condition OFFSET = TC = SGND IFB = 0 to 75A ILOAD=0; IOFFSET =100A; TC = SGND ILOAD=0; IOFFSET =100mA; TC Enabled IOFFSET VOFFSET OFFSET pin Current Range OFFSET pin Voltage IOFFSET = 0 to 250A Min. -3.5 -90 90 0 Typ. -100 100 1.240 Max. +3.5 -110 110 250 Unit A A A A V THERMAL SENSOR VTC TC Voltage at Tamb = 27C RTC=100k RTC=50k RTC=5k 0.612 0.593 0.530 0.645 0.625 0.558 0.677 0.656 0.586 V GATE DRIVERS tRISE HGATE IHGATEx RHGATEx tRISE LGATE ILGATEx RLGATEx High Side Rise Time High Side Source Current High Side Sink Resistance Low Side Rise Time Low Side Source Current Low Side Sink Resistance BOOTx-PHASEx=10V; CHGATEx to PHASEx=3.3nF BOOTx-PHASEx=10V BOOTx-PHASEx=12V; VCCDRx=10V; CLGATEx to PGNDx=5.6nF VCCDRx=10V VCCDRx=12V 0.7 1.5 15 2 2 30 1.8 1.1 1.5 2.5 55 30 ns A ns A PROTECTIONS AND SS_END VSS_ENDL ISS_ENDH UVP OVP SS_END Voltage Low SS_END Leakage Under Voltage Trip Over Voltage Threshold Over Voltage Threshold Preliminary OVP Turn-ON Threshold PreOVP Threshold PreOVP Hysteresis ISS_END = -4mA VSS_END = 5V VSEN Falling VSEN Rising; OVP = OPEN VSEN Rising; OVP = 90k* to SGND VCC = VCCDRx Rising FBR Rising 55 112 1.54 60 115 1.64 4 1.8 350 0.4 1 65 118 1.74 V A % % V V V mV 8/38 L6711 Table 5. Voltage IDentification (VID) Codes. VID_SEL = OPEN (VRD 10.x DAC with -19mV auto-offset) VID5 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 VID4 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 VID3 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 VID2 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 VID1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 VID0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 Output Voltage (V) (*) 0.8185 0.8310 0.8435 0.8560 0.8685 0.8810 0.8935 0.9060 0.9185 0.9310 0.9435 0.9560 0.9685 0.9810 0.9935 1.0060 1.0185 1.0310 1.0435 1.0560 1.0685 OFF OFF 1.0810 1.0935 1.1060 1.1185 1.1310 1.1435 1.1560 1.1685 1.1810 VID5 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 VID4 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 VID3 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 VID2 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 VID1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 VID0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 Output Voltage (V) (*) 1.1935 1.2060 1.2185 1.2310 1.2435 1.2560 1.2685 1.2810 1.2935 1.3060 1.3185 1.3310 1.3435 1.3560 1.3685 1.3810 1.3935 1.4060 1.4185 1.4310 1.4435 1.4560 1.4685 1.4810 1.4935 1.5060 1.5185 1.5310 1.5435 1.5560 1.5685 1.5810 (*) Since the VIDx pins program the maximum output voltage, according to VRD 10.x specs, the device automatically regulates to a voltage 19mV lower avoiding use of any external component to lower the regulated voltage. This improves the system tolerance performance since the reference already offset is trimmed during production within 0.5%. 9/38 L6711 Table 6. Voltage IDentification (VID) Codes. VID_SEL = SGND (Hammer DAC) HAMMER DAC VID5 VID4 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 VID3 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 VID2 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 VID1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 VID0 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 Output Voltage (V) 1.550 1.525 1.500 1.475 1.450 1.425 1.400 1.375 1.350 1.325 1.300 1.275 1.250 1.225 1.200 1.175 1.150 1.125 1.100 1.075 1.050 1.025 1.000 0.975 0.950 0.925 0.900 0.875 0.850 0.825 0.800 OFF 0 VID5 VID4 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 HAMMER DAC +25mV VID3 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 VID2 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 VID1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 VID0 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 Output Voltage (V) 1.575 1.550 1.525 1.500 1.475 1.450 1.425 1.400 1.375 1.350 1.325 1.300 1.275 1.250 1.225 1.200 1.175 1.150 1.125 1.100 1.075 1.050 1.025 1.000 0.975 0.950 0.925 0.900 0.875 0.850 0.825 OFF 10/38 L6711 Figure 4. PRINCIPLE SCHEMATIC 1 Low Side Mosfet Current Sense VIN LIN GNDIN 1 41 38 CIN VCCDR1 VCCDR2 VCCDR3 BOOT1 45 4 VCC UGATE1 46 HS1 L1 5,31 22 6 PHASE1 SGND OSC/FAULT OFFSET LGATE1 47 2 LS1 PGND1 CS1- 3 14 Rg 20 TC CS1+ 15 Rg 21 9 CS_SEL OVP BOOT2 44 UGATE2 43 HS2 L2 Vcc_core L6711 VID5 VID4 VID3 VID2 VID1 VID0 VID_SEL OUTEN 23 28 27 26 25 24 29 13 VID5 VID4 VID3 VID2 VID1 VID0 VID_SEL OUTEN PHASE2 42 LGATE2 40 LS2 COUT LOAD PGND2 CS2- 39 16 Rg CS2+ BOOT3 17 34 Rg 7 COMP UGATE3 33 HS3 L3 CF RF PHASE3 8 FB 32 LGATE3 36 LS3 RFB 10 PGND3 CS3VSEN CS3+ FBR FBG SS_END N.C. 37 N.C. 48 35 18 Rg SS_END 19 Rg 11 12 30 L6711 REF. SCH. (MOSFET) 11/38 L6711 Figure 5. PRINCIPLE SCHEMATIC 2 Inductor Current Sense VIN LIN GNDIN 1 41 38 CIN VCCDR1 VCCDR2 VCCDR3 BOOT1 45 4 VCC UGATE1 46 HS1 L1 5,31 22 6 PHASE1 SGND OSC/FAULT OFFSET LGATE1 47 2 LS1 Rg(RC) PGND1 CS1- 3 Cg 14 Rg 20 TC CS1+ 15 Rg(a) 21 9 CS_SEL OVP BOOT2 44 UGATE2 43 HS2 L2 Vcc_core L6711 VID5 VID4 VID3 VID2 VID1 VID0 VID_SEL OUTEN 23 28 27 26 25 24 29 13 VID5 VID4 VID3 VID2 VID1 VID0 VID_SEL OUTEN PHASE2 42 LGATE2 40 LS2 Rg(RC) COUT LOAD PGND2 CS2- 39 Cg 16 Rg CS2+ BOOT3 17 34 Rg(a) 7 COMP UGATE3 33 HS3 L3 CF RF PHASE3 8 FB 32 LGATE3 36 LS3 Rg(RC) RFB 10 PGND3 CS3VSEN CS3+ FBR FBG SS_END N.C. 37 N.C. 48 35 Cg 18 R SS_END 19 Rg(a) 11 12 30 L6711 REF. SCH. (INDUCTOR) 12/38 L6711 4 DEVICE DESCRIPTION The device is a three phase PWM controller with embedded high current drivers that provides complete control logic and protections for a high performance step-down DC-DC voltage regulator optimized for advanced microprocessor power supply. Multi phase buck is the simplest and most cost-effective topology employable to satisfy the increasing current demand of newer microprocessors and modern high current DC/DC converters and POLs. It allows distributing equally load and power between the phases using smaller, cheaper and most common external power mosfets and inductors. Moreover, thanks to the 120 of phase shift between each phase, the input and output capacitor count results in being reduced. Phase interleaving causes in fact input rms current and output ripple voltage reduction and show an effective output switching frequency increase: the 150kHz free-running frequency per phase, externally adjustable through a resistor, results tripled on the output. The controller includes multiple DACs, selectable through an apposite pin (VID_SEL), allowing compatibility with both VRD 10.x and Hammer specifications, also performing D-VID transitions accordingly. The output voltage can be precisely selected, programming the VID and VID_SEL pins, from 0.8185V to 1.5810V with 12.5mV binary steps (VRD 10.x compliant mode - 6 BIT with -19mV offset already programmed during production) or from 0.800V to 1.550V with 25mV steps (VRM Hammer compliant mode - 5 BIT, VID5 programs a 25mV positive offset in this case), with a maximum tolerance on the output regulated voltage of 0.5% (0.6% for Hammer) over temperature and line voltage variations. The device permits easy and flexible system design by allowing current reading across either inductor or low side mosfet in fully differential mode simply selecting the desired way through the CS_SEL pin. In both cases, also a sense resistor in series to the related element can be considered to improve reading precision. The current information read corrects the PWM output in order to equalize the average current carried by each phase limiting the error at 3% over static and dynamic conditions unless considering the sensing element spread. The device provides a programmable Over-Voltage protection to protect the load from dangerous over stress and can be externally set to a fixed voltage through an apposite resistor or it can be set internally with a fixed percentage, latching immediately by turning ON the lower driver and driving high the FAULT pin. Furthermore, preliminary OVP protection also allows the device to protect load from dangerous OVP when VCC is not above the UVLO threshold. Over-Current protection provided, with an OC threshold for each phase, causes the device to enter in constant current mode until the latched UVP. Depending on the reading mode selected, the device keeps constant the peak (inductor sensing) or the valley (LS sensing) of the inductor current ripple. The device drives high the FAULT pin after each latching event: to recover it is enough to cycle VCC or the OUTEN pin. A compact 7x7mm body TQFP48 package with exposed thermal pad allows dissipating the power to drive the external mosfet through the system board. 5 CURRENT READING AND CURRENT SHARING CONTROL LOOP The device embeds a flexible, fully-differential current sense circuitry that is able to read across both low side or inductor parasitic resistance or across a sense resistor placed in series to that element. The fullydifferential current reading rejects noise and allows placing sensing element in different locations without affecting the measurement's accuracy. The kind of sense element can be simply chosen through the CS_SEL pin: setting this pin free, the LS mosfet is used while shorting it to SGND, the inductor will be used instead. Details about connections are shown in Figure 6. The high bandwidth current sharing control loop allows current balance even during load transients: a current reference equal to the average of the read current (IAVG) is internally built and the error between the read current and this reference is converted to a voltage that with a proper gain is used to adjust the duty cycle whose dominant value is set by the voltage error amplifier. 13/38 L6711 Figure 6. Current Reading Connections selectable through CS_SEL pin. CS_SEL LGATEx Rg CSx- CS_SEL PHASEx IPHASE L Rg(RC) Cg Rg(a) RL OUT CSx+ IPHASE Rg CSx+ CSx- Rg LS Mosfet Current Sense Inductor Current Sense 5.1 LOW SIDE Current Reading Leaving CS_SEL pin OPEN, the current flowing trough each phase is read using the voltage drop across the low side mosfets RdsON or across a sense resistor in its series and it is internally converted into a current. The transconductance ratio is issued by the external resistor Rg placed outside the chip between CSx- and CSx+ pins toward the reading points (see Figure 7 right). The proprietary current sense circuit tracks the current information for a time TTRACK = TSW/3 (TSW = 1/FSW) centered in the middle of the lowside mosfet conduction time (OFF Time, see Figure 7 left) and holds the tracked information during the rest of the period. This device sources a constant 50A current from the CSx+ pin: the current reading circuitry uses this pin as a reference and the reaction keeps the CSx- pin to this voltage during the reading time (an internal clamp keeps CSx+ and CSx- at the same voltage sinking from the CSx- pin the necessary current during the hold time; this is needed when LS mosfet RdsON sense is implemented to avoid absolute maximum rating overcome on CSx- pin). The current that flows from the CSx- pin is then given by the following equation (See Figure 7 - right): R dsON R dsON I CSx- = 50A + ----------------- IPHASEx = 50A + I INFOx where I INFOx = ----------------- I PHASEx Rg Rg RdsON is the on resistance of the low side mosfet and Rg is the transconductance resistor used between CSx- and CSx+ pins toward the reading points; IPHASEx is the current carried by the relative phase and IINFOx is the current information signal reproduced internally. 50A offset allows negative current reading, enabling the device to check for dangerous returning current between the phases assuring the complete current equalization. From the current information of each phase, information about the total current delivered (IDROOP = IINFO1 + IINFO2 + IINFO3) and the average current for each phase (IAVG = (IINFO1 + IINFO2 + IINFO3)/3 ) is taken. IINFOX is then compared to IAVG to give the correction to the PWMx output in order to equalize the current carried by the three phases. Figure 7. Current reading across LS mosfet: timing (left) and circuit (right) for each phase. IPHASEx ILSx LGATEx Rg CSx- ICSx- IPHASEx Rg IINFOx 50A CSx+ TTRACK TSW 14/38 L6711 5.2 INDUCTOR CURRENT READING Shorting CS_SEL pin to SGND, the current flowing trough each phase is read using the voltage drop across the output inductor or across a sense resistor (RSENSE) in its series and internally converted into a current. The transconductance ratio is issued by the external resistor Rg placed outside the chip between CSx- and CSx+ pins toward the reading points (see Figure 6 right). The current sense circuit always tracks the current sensed and still sources a constant 50A current from the CSx+ pin: this pin is used as a reference keeping the CSx- pin to this voltage. To correctly reproduce the inductor current an R-C filtering network must be introduced in parallel to the sensing element. The current that flows from the CSx- pin is then given by the following equation (See Figure 8): L 1 + s -----RL RL ICSx- = 50A + ------ ------------------------------------------------ IPHASEx R g 1 + s R g ( RC ) Cg Where IPHASEx is the current carried by the relative phase. Considering now to match the time constant between the inductor and the R-C filter applied (Time constant mismatches cause the introduction of poles into the current reading network causing instability. Moreover, it is also important for the load transient response and to let the system show resistive equivalent output impedance), it results: RL RL L ------ = R g ( RC ) Cg ICSx- = 50A + ------- IPHASEx = 50A + I INFOx where I INFOx = IPHASEx ------Rg Rg RL Where IINFOx is the current information reproduced internally. 50A offset allows negative current reading, enabling the device to check for dangerous returning current between the phases assuring the complete current equalization. From the current information of each phase, information about the total current delivered (IDROOP = IINFO1 +IINFO2 + IINFO3) and the average current for each phase (IAVG = (IINFO1 + IINFO2 + IINFO3)/3) is taken. IINFOX is then compared to IAVG to give the correction to the PWM output in order to equalize the current carried by the three phases. Since Rg is designed considering the OC protection, to allow further flexibility in the system design, the resistor in series to CSx+ can be split in two resistors as shown in Figure 8. Figure 8. Inductor Current Sense CS_SEL PHASEx IPHASE L Rg(RC) Cg Rg(a) Rg RL OUT Design Equations: LR g ( RC) = -----------------RL Cg R g ( RC) + R g ( a ) = Rg CSx+ 50A ICSx- CSx- 15/38 L6711 6 DAC SELECTION The device embeds a selectable DAC that allows the output voltage to have a tolerance of 0.5% (0.6% for Hammer DAC) recovering from offsets and manufacturing variations. The VID_SEL pin selects the DAC table used to program the reference for the regulation as shown in Table 7. Table 7. DAC Selection VID_SEL Selected DAC VRM / VRD 10.x DAC. Output voltage ranges from 0.8185V to 1.5810V with 12.5mV steps (See Table 5). Since the VIDx pins program the maximum output voltage, according to VRD 10.x specs, the device automatically regulates with -19mV offset avoiding use of any external component to lower the regulated voltage. Since the -19mV offset is programmed during the production stage, no further error is introduced to generate the offset since it is automatically recovered during the trimming stage. VID5 OPEN SGND SGND Hammer DAC Output voltage ranges from 0.800V to 1.550V with 25mV steps (See Table 6). Output voltage ranges from 0.825V to 1.575V with 25mV steps (See Table 6). Since the +25mV offset is programmed during the production stage, no further error is introduced to generate the offset since it is automatically recovered during the trimming stage. OPEN VID pins are inputs of an internal DAC that is realized by means of a series of resistors providing a partition of the internal voltage reference. The VID code drives a multiplexer that selects a voltage on a precise point of the divider. The DAC output is delivered to an amplifier obtaining the voltage reference (i.e. the set-point of the error amplifier, VPROG). Internal pull-ups are provided (realized with a 5A current generator up to 3V Typ); in this way, to program a logic "1" it is enough to leave the pin floating, while to program a logic "0" it is enough to short the pin to SGND. Programming the "11111x" code (NOCPU, VID5 is irrelevant), the device shuts down: all mosfets are turned OFF and SS_END is shorted to SGND. Removing the code causes the device to restart. The voltage identification (VID) pin configuration also sets the Over / Under Voltage protection (OVP/UVP) thresholds. 7 REMOTE VOLTAGE SENSE The device embeds a Remote Sense Buffer to sense remotely the regulated voltage without any additional external components. In this way, the output voltage programmed is regulated between the remote buffer inputs compensating motherboard or connector losses. The very low-offset amplifier senses the output voltage remotely through the pins FBR and FBG (FBR is for the regulated voltage sense while FBG is for the ground sense) and reports this voltage internally at VSEN pin with unity gain eliminating the errors. Keeping the FBR and FBG traces parallel and guarded by a power plane results in common mode coupling for any picked-up noise. If remote sense is not required, it is enough connecting the resistor RFB directly to the regulated voltage: VSEN becomes not connected and still senses the output voltage through the remote buffer. In this case the FBG and FBR pins must be connected anyway to the regulated voltage (See Figure 9). 7.1 Warning: The remote buffer is included in the trimming chain in order to achieve 0.5% accuracy (0.6% for the Hammer DAC) on the output voltage when the RB is used: eliminating it from the control loop causes the regulation error to be increased by the RB offset worsening the device performances! 16/38 L6711 Figure 9. Remote Buffer Connections 64k 64k VID 64k 64k 64k VID 64k 64k 64k FBR FB RF RFB COMP CF VSEN FBR FBG FB RF COMP CF VSEN FBG To Vcore (Remote Sense) RFB To Vcore RB used (up to 0.5% Accuracy) RB Not Used (Precision worsened) 8 VOLTAGE POSITIONING Output voltage positioning is performed by selecting the reference DAC and by programming the different contributors to the IFB current (see Figure 10). This current, sourced from the FB pin, causes the output voltage to vary according to the external RFB resistor: this allows programming precise output voltage variations depending on the sensed current (Droop Function) as well as offsets for the regulation. The three contributors to the IFB current value are: Droop Function (green); Offset (red); Integrated Temperature Compensation (fuchsia). Moreover, the embedded Remote Buffer allows to precisely programming the output voltage offsets and variations by recovering the voltage drops across distribution lines. The output voltage is then driven by the following relationship (IOFFSET sign depends on TC setting): V OUT = VID - R FB IFB = VID - R FB ( I DROOP I OFFSET - I TC ) Figure 10. Voltage Positioning and Droop Function ITC ESR DROP 64k IOFFSET IDROOP VID 64k IFB 64k VMAX VNOM VMIN RESPONSE WITHOUT DROOP RESPONSE WITH DROOP 64k FB RF RFB COMP CF VSEN FBR FBG To Vcore (Remote Sense) 8.1 DROOP FUNCTION Droop function allows the device to satisfy the requirements of high performance microprocessors, reducing the size and the cost of the output capacitor. This method "recovers" part of the drop due to the output capacitor ESR in the load transient, introducing a dependence of the output voltage on the load current: a static error proportional to the output current causes the output voltage to vary according to the sensed current. As shown in figure 4-right, the ESR drop is present in any case, but using the droop function the total deviation of the output voltage is minimized. 17/38 L6711 The information about the total current delivered (IDROOP) is sourced from the FB pin (see Figure 10): connecting a resistor between this pin and VSEN (i.e. the output voltage), the total current information flows only in this resistor because the compensation network between FB and COMP has always a capacitor in series (CF, see Figure 10). The voltage regulated is then equal to: VOUT = VID - R FB IDROOP Where VID is the reference programmed through VIDx and VID_SEL (Only the IDROOP contribute to IFB has been considered). Since IDROOP depends on the current information about the three phases, the output characteristic vs. load current is given by: R SENSE V OUT = VID - R FB I DROOP = VID - R FB --------------------- IOUT = VID - RDROOP I OUT Rg Where RSENSE is the chosen sensing element resistance (Inductor DCR or LS RdsON), IOUT is the output current of the system and RDROOP is its equivalent output resistance (The whole power supply can be then represented by a "real" voltage generator with a voltage value of VID and an equivalent series resistance RDROOP). RFB resistor can be also designed according to the RDROOP specifications as follow: Rg R FB = R DROOP --------------------R SENSE 8.2 OFFSET The OFFSET pin allows programming a positive or a negative offset (VOS) for the output voltage. When the Integrated Thermal Sensor is disabled (TC = SGND) a resistor ROFFSET connected vs. SGND increases the output voltage: since the pin is internally fixed at 1.240V, the current programmed by the resistor ROFFSET is mirrored and then properly subtracted from the IFB current (see Figure 11) as follow (Only the IOFFSET contribute to IFB has been considered): 1.240V V OUT = VID + R FB I OFFSET = VID + R FB ------------------------ = VID + V OS R OFFSET Figure 11. Voltage Positioning with Offset 64k IOFFSET IOFFSET 1.240V 1:1 IFB IDROOP VID 64k 64k 64k TC OFFSET ROFFSET FB RF RFB COMP CF VSEN FBR FBG To Vcore (Remote Sense) The device will add the programmed offset VOS to the output programmed voltage (considering now also the droop effect) subtracting the relative offset current from the feedback current IFB: VOUT = VID - R FB IFB = VID - R FB ( I DROOP - IOFFSET ) = VID + R FB IOFFSET - R DROOP I OUT Offset resistor can be designed by considering the following relationship (RFB is fixed by the Droop effect): 18/38 L6711 1.240V R OFFSET = ------------------ R FB VOS Offset automatically given by the DAC selection or by VID5 when VID_SEL=SGND differs from the offset implemented through the OFFSET pin: the built-in feature is trimmed in production and assures 0.5% error (0.6% for the Hammer DAC) over load and line variations while implementing the same offset through the OFFSET pin causes additional errors to be considered in the total output voltage precision. When the Integrated Thermal Sensor is enabled (see Figure 12 and following section), the pin programs, in the same way as before, a negative offset. This is to compensate the positive native offset introduced by the ITS. The effect of the programmed offset on the output voltage results (IOFFSET is now added to IFB and no more subtracted as before): 1.240 V OUT = VID - R FB IOFFSET = VID - R FB ------------------------ = VID - V OS R OFFSET Offset resistor is designed to compensate the ITS native offset as described in the following section. The Offset function can be disabled by shorting the pin to SGND. Figure 12. Voltage Positioning with Integrated Thermal Sensor 1:1 ITC 64k VTC=A+B*(TJ-25) IOFFSET IDROOP VID 64k IOFFET 1.240V ITC 1:1 IFB 64k TC RTC OFFSET ROFFSET FB RF COMP CF VSEN FBR 64k FBG To Vcore (Remote Sense) RFB 8.3 INTEGRATED THERMAL SENSOR Current sense elements have non-negligible temperature variations: considering either inductor or LS mosfet sense, the sensing elements modify proportionally to varying temperature. As a consequence, the sensed current is subjected to a measurement error that causes the regulated voltage to vary accordingly. To recover from this temperature related error, a temperature compensation circuit is integrated into the controller: the internal temperature is sensed and the droop current is corrected (according to a scaling external resistor RTC) in order to keep constant the regulated voltage. The ITS circuit subtracts from the IFB current a current proportional to the sensed temperature as follow (see Figure 12, Only the IDROOP and ITC contributes to IFB have been considered): R SENSE ( T MOS ) V OUT ( T,IOUT ) = VID - R FB ------------------------------------------ IOUT - I TC ( T J ) Rg 1where I TC ( T J ) = ---------- [ A + B ( T J - 25 ) ] R TC where A and B are positive constants depending on the value of the external resistor RTC (see Figure 13), TJ is the device junction temperature and TMOS is the mosfet (or the used sensing element) temperature. The resistor RTC can be designed in order to zero the temperature influence on the output voltage at a fixed current as follow: 19/38 L6711 R SENSE ( T MOS - 25 ) R FB - R FB ------------------------------------------------------------------- IOUT + ---------- B ( T J - 25 ) = 0 Rg R TC obtaining the following relationship: B kT Rg R TC = --------------------- -------------------R SENSE I OUT where RSENSE is the sensing element resistance value (at TMOS = 25C), B is the constant obtainable from Figure 13, kT is the Temperature Coupling Coefficient between the sensing element and the Controller (it results KT = (TJ-25)/(TMOS-25)) and is the Temperature Coefficient of the sensing element. Since RTC depends from the constant B depending in turn from RTC, an iterative process is required to properly design the RTC value. As a consequence of the nature of the thermal sensor, a negative offset is needed to compensate the native offset introduced by the ITS at a referenced temperature Tref and it is obtainable by connecting a ROFFSET resistor between the OFFSET pin and SGND as follow: A + B ( T ref - 25 ) I OFFSET = - I TC ( T ref ) = --------------------------------------------R TC To disable this function, short the pin to SGND. Figure 13. Integrated Thermal Sensor Constant vs. External resistor RTC 650 625 600 575 550 525 500 475 450 0 10 20 30 40 50 60 70 80 90 100 1.70 1.60 1.50 0 10 20 30 40 50 60 70 80 90 100 2.00 1.90 1.80 A [mV] vs. RTC [K] B [mV/C] vs. RTC [K] 9 DYNAMIC VID TRANSITION The device is able to manage Dynamic VID Code changes that allow Output Voltage modification during normal device operation. OVP and UVP signals are masked during every VID transition and they are re-activated after the transition finishes. When changing dynamically the regulated voltage (D-VID), the system needs to charge or discharge the output capacitor accordingly. This means that an extra-current ID-VID needs to be delivered, especially when increasing the output regulated voltage and it must be considered when setting the over current threshold. This current result: C OUT dV OUT I D - VID = -------------------------------------dT VID where dVOUT is the selected DAC LSB (12.5mV for VRD10.x or 25mV for Hammer DAC) and TVID is the time interval between each LSB transition. 20/38 L6711 Overcoming the OC threshold during the dynamic VID causes the device to enter the constant current limitation slowing down the output voltage dV/dt also causing the failure in the D-VID test. The way in which the device modifies the reference depends on the VID_SEL status and then on the kind of DAC selected. 9.1 VID_SEL = OPEN (see Figure 14). Selecting the VRD10.x DAC, the device checks for VID code modifications on the rising edge of a clock that is three times the switching frequency of each phase and waits for a confirmation on the following falling edge. Once the new code is stable, on the next rising edge, the reference starts stepping up or down in LSB increments (12.5mV) every clock cycle (still 3*FSW) until the new VID code is reached. During the transition, VID code changes are ignored; the device re-starts monitoring VID after the transition has finished on the next rising edge available. 9.1.1 Warning: if the new VID code is more than 1 LSB higher than the previous, the device will execute the transition stepping the reference with a frequency equal to 3*FSW until the new code has reached: for this reason it is recommended to carefully control the VID change rate in order to carefully control the slope of the output voltage variation. Figure 14. Dynamic VID Transition, VRD10.x DAC Ref Moved (2) Ref Moved (3) Ref Moved (4) VID Sampled VID Stable Ref Moved (1) VID Sampled VID Stable Ref Moved (1) VID Sampled VID Stable Ref Moved (1) VID Sampled VID Stable Ref Moved (1) VID Sampled VID Stable Ref Moved (1) VID Sampled VID Sampled VID Sampled VID Sampled VID Sampled VID Sampled VID Sampled VID Sampled VID Sampled VID Sampled VID Sampled VID Clock VID Sampled VID [0,5] t Int. Reference Tsw/3 t Tsw Vout TVID t x 4 Step VID Transition VRD10.x DAC 4 x 1 Step VID Transition - VRD10.x DAC t 21/38 L6711 Figure 15. Dynamic VID Transition, Hammer DAC Ref Moved (1) Ref Moved (3) Ref Moved (2) Ref Moved (4) VID Sampled VID Sampled VID Changed VID Sampled VID Sampled VID Stable VID Clock VID [0,5] t Int. Reference t Tsw Vout t 4 Step VID Transition - Hammer DAC t 9.2 VID_SEL = GND (see Figure 15). Selecting the HAMMER DAC, the device checks for VID code modifications on the rising edge of a clock that is the same frequency of each phase and waits for a confirmation on the following falling edge. Once the new code is stable, on the next rising edge the reference starts stepping up or down in LSB increments (25mV) every clock cycle until the new VID code is reached. During the transition, VID code changes are ignored; the device re-starts monitoring VID after the transition has finished on the next rising edge. If the new VID code is more than 1 bit higher than the previous, the device will execute the transition stepping the reference every switching cycle until the new code has reached 10 ENABLE AND DISABLE The device has three different supplies: VCC pin to supply the internal control logic, VCCDRx to supply the low side drivers and BOOTx to supply the high side drivers. If the voltage at pins VCC and VCCDRx are not above the turn on thresholds specified in the Electrical Characteristics, the device is shut down: all drivers keep the mosfets off to show high impedance to the load. Once the device is correctly supplied, proper operation is assured but the device can be controlled in different ways: 10.1 OUTEN pin It can be used to control the power sequencing in complex systems. Setting the pin free, the device implements a soft start up to the programmed voltage. Shorting the pin to SGND, it resets the device (SS_END is shorted to SGND in this condition and protections are disabled except pre-OVP) from any latched condition and also disables the device keeping all the mosfet turned off to show high impedance to the load. It can be then cycled to recover from any latched condition such as OVP and UVP. 10.2 NOCPU (VID [0;5]=11111x) In this condition (VID5 state is irrelevant) the device is disabled and keeps all the mosfet turned off to show high impedance to the load. Nevertheless, it waits for any VID code transition to power up implementing a soft start. During this condition, SS_END pin is shorted to SGND. 22/38 L6711 11 SOFT START During soft start, a ramp is generated increasing the loop reference from 0V to the final value programmed by VID in 2048 clock periods as shown in Figure 16. Once the soft start begins, the reference is increased: upper and lower MOS begin to switch and the output voltage starts to increase with closed loop regulation. At the end of the digital soft start, the SS_END signal is then driven high. The Under Voltage comparator is enabled when the reference voltage reaches 0.6V while Over Voltage Comparator is always enabled during soft start with a threshold equal to the 115% of the programmed reference or the threshold programmed by ROVP (see relevant section). Figure 16. Soft Start VCC=VCCDR Turn ON threshold VLGATEx t VOUT t SS_END t 2048 Clock Cycles t (CH1=VOUT; CH2=LGATEx; CH3=SS_END) 12 OUTPUT VOLTAGE MONITOR AND PROTECTIONS The device monitors through pin VSEN the regulated voltage in order to manage the OVP / UVP conditions. 12.1 UVP Protection If the output voltage monitored by VSEN drops below the 60% of the reference voltage for more than one clock period, the device turns off all mosfets and the OSC/FAULT is driven high (5V). The condition is latched; to recover it is required to cycle Vcc or the OUTEN pin. 12.2 Programmable OVP Protection Once VCC crosses the turn-ON threshold and the device is enabled (OUTEN = 1), the device provides a programmable Over Voltage protection; when the voltage sensed overcomes the programmed threshold, the controller permanently switches on all the low-side mosfets and switches off all the high-side mosfets in order to protect the load. The OSC/ FAULT pin is driven high (5V) and power supply or OUTEN pin cycling is required to restart operations. The OVP threshold is programmed through the OVP pin: leaving the pin floating, it is internally pulled-up and the threshold is set at 115% (Typ.) of the programmed output voltage. Connecting the OVP pin to SGND through a resistor ROVP, the OVP threshold becomes a fixed voltage as follow: OVPTH = 1.455 * ROVP * 12.5 12.3 Preliminary OVP Protection (Pre-OVP) While VCC pin is under the turn-ON threshold, a preliminary-OVP protection turns on the low side mosfets as long as the FBR pin voltage is greater than 1.8V. This protection is enabled when VCC stays within the 23/38 L6711 device turn-on threshold and the PreOVP turn on threshold and depends also on the OUTEN pin status as detailed in Figure 17 - left. A simple way to provide protection to the output in all conditions when the device is OFF (then avoiding the unprotected red region in Figure 17) consists in supplying the controller through the 5VSB bus as shown in Figure 17 - right. Both Over Voltage and Under Voltage are active also during soft start (see the relevant section). Figure 17. Output Voltage Protections and typical principle connections to assure complete protection. +5V Vcc Vcc TurnON (OUTEN = 0) Preliminary OVP FBR Monitored (OUTEN = 1) Programmable OVP VSEN Monitored SB +12V VCC Preliminary OVP Enabled FBR Monitored PreOVP TurnON No Protection Provided VCCDR1 VCCDR2 VCCDR3 13 OVERCURRENT PROTECTION Depending on the current reading method selected, the device limits the peak or the bottom of the inductor current entering in constant current until setting UVP as below explained. The Over Current threshold has to be programmed, by designing the Rg resistors, to a safe value, in order to be sure that the device doesn't enter OCP during normal operation of the device. This value must take into consideration also the extra current needed during the Dynamic VID Transition ID-VID and, since the device reads across mosfets RdsON or inductor DCR, the process spread and temperature variations of these sensing elements. Moreover, since also the internal threshold spreads, the Rg design must consider its minimum value IOCTH(min) as follow: I OCPx ( max ) R SENSE ( max Rg = ----------------------------------------------------------------------) IOCTH ( min ) where IOCPx is the current measured by the current reading circuitry when the device enters Quasi Constant Current (LS Mosfet Sense) or Constant Current (Inductor Sense), IOCPx must be calculated starting from the corresponding output current value IOUT(OCP) as follow (ID-VID must also be considered when D-VID are implemented): I OCPx = I OUT( OCP ) Ipp I D-VID --------------------------- - ---------- + -------------- Low Side Mosfet Sense 3 2 3 I OUT( OCP I pp I D-VID ---------------------------) + ---------- + -------------- Inductor DCR Sense 3 2 3 where IOUT(OCP) is still the output current value at which the device enters Quasi Constant Current (LS Mosfet Sense) or Constant Current (Inductor Sense),IPP is the inductor current ripple in each phase and ID-VID is the additional current required by D-VID (when applicable). In particular, since the device limits the peak or the valley of the inductor current (according to CS_SEL status), the ripple entity, when not negligible, impacts on the real OC threshold value and must be considered. 24/38 L6711 13.1 LS MOSFET SENSE (CS_SEL=OPEN) OVERCURRENT The device detects an Over Current condition for each phase when the current information IINFOx overcomes the fixed threshold of IOCTH (35A Typ). When this happens, the device keeps the relative LS mosfet on, skipping clock cycles, until the threshold is crossed back and IINFOx results being lower than the IOCTH threshold. This implies that the device limits the bottom of each inductor current ripple. After exiting the OC condition, the LS mosfet is turned off and the HS is turned on with a duty cycle driven by the PWM comparator. Keeping the LS on, skipping clock cycles, causes the on-time subsequent to the exit from the OC condition to increase. Considering now that the device, with this kind of current sense, has maximum on-time dependence with the delivered current given by the following relationship: R SENSE 0.80 T SW I DROOP = 0A T ON,MAX = ( 0.80 - IDROOP 3.8k ) TSW = 0.80 - --------------------- I OU T 3.8k T SW = Rg 0.40 T SW I DROOP = 105A Where IOUT is the output current (IOUT = *IPHASEx) and TSW is the switching period (TSW =1/FSW). This linear dependence has a value at zero load of 0.80 *TSW and at maximum current of 0.40 *TSW typical and results in two different over current behaviors of the device: 13.2 TON Limited Output Voltage. This happens when the maximum ON time is reached before that the current in each phase reaches IOCPx (IINFOx TON Limited Output characteristic Resulting Output characteristic Desired Output characteristic and UVP threshold 0.40*VIN 0.40*VIN IOCP=3*IOCPx (IDROOP=105A) IOUT IOCP=3*IOCPx (IDROOP=105A) IOUT Maximum output Voltage b) TON Limited Output Voltage 13.2.1Constant Current Operation This happens when the on-time limitation is reached after the valley current in each phase reaches IOCPx (IINFOx > IOCTH). The device enters in Quasi-Constant-Current operation: the low-side mosfets stays ON until the current read becomes lower than IOCPx (IINFOx < IOCTH) skipping clock cycles. The high side mosfet can be then turned ON with a TON imposed by the control loop after the LS turn-off and the device works in the usual way until another OCP event is detected. This means that the average current delivered can slightly increase in Quasi-Constant-Current operation 25/38 L6711 since the current ripple increases. In fact, the ON time increases due to the OFF time rise because of the current has to reach the IOCPx bottom. The worst-case condition is when the ON time reaches its maximum value. When this happens, the device works in Constant Current and the output voltage decrease as the load increase. Crossing the UVP threshold causes the device to latch driving high the OSC pin (Figure 19 shows this working condition). It can be observed that the peak current (Ipeak) is greater than IOCPx but it can be determined as follow: V IN - Vout MIN V IN - Vout MIN I PEAK = I OCPx + -------------------------------------- Ton MAX = I OCPx + -------------------------------------- 0.40 T SW L L Where VoutMIN is the UVP threshold, (inductor saturation must be considered). When that threshold is crossed, all mosfets are turned off, the FAULT pin is driven high and the device stops working. Cycle the power supply or the OUTEN pin to restart operation. The maximum average current during the Constant-Current behavior results: Ipeak - I OCPx IMAX,TOT = 3 IMAX = 3 IOCPx + ------------------------------------- 2 In this particular situation, the switching frequency for each phase results reduced. The ON time is the maximum allowed (TonMAX) while the OFF time depends on the application: Ipeak - IOCPx T OFF = L ------------------------------------V out 1 = ---------------------------------------T onMax + T OFF The transconductance resistor Rg can be designed considering that the device limits the bottom of the inductor current ripple and also considering the additional current delivered during the quasi-constant-current behavior as previously described in the worst case conditions. Moreover, when designing D-VID compatible systems, the additional current due to the output filter charge during dynamic VID transitions must be considered. IOUT ( OCP ) I PP ID-VID IOCPx ( max ) R SENSE ( max Rg = ----------------------------------------------------------------------) where I OCPx = --------------------------- - ----------- + -------------IOCTH ( min ) 2 3 3 Figure 19. Constant Current operation Ipeak IMAX IOCPx Vout Droop effect UVP TonMAX TonMAX IMAX,TOT Iout 3*IOCPx (IDROOP=105A) a) Maximum current for each phase b) Output Characteristic 13.3 INDUCTOR SENSE (CS_SEL = SGND) OVER CURRENT The device detects an over current when the IINFOx overcome the fixed threshold IOCTH. Since the device always senses the current across the inductor, the IOCTH crossing will happen during the HS conduction time: as a consequence of OCP detection, the device will turn OFF the HS mosfet and turns ON the LS 26/38 L6711 mosfet of that phase until IINFOx re-cross the threshold or until the next clock cycle. This implies that the device limits the peak of the inductor current. In any case, the inductor current won't overcome the IOCPx value and this will represent the maximum peak value to consider in the OC design. The device works in Constant-Current, and the output voltage decreases as the load increase, until the output voltage reaches the UVP threshold. When this threshold is crossed, all mosfets are turned off, the FAULT pin is driven high and the device stops working. Cycle the power supply or the OUTEN pin to restart operation. The transconductance resistor Rg can be designed considering that the device limits the inductor current ripple peak. Moreover, when designing D-VID systems, the additional current due to the output filter charge during dynamic VID transitions must be considered. I OUT ( OCP I PP ID-VID IOCPx ( max ) R SENSE ( max Rg = ----------------------------------------------------------------------) where I OCPx = ---------------------------) + ----------- + -------------I OCTH ( min ) 2 3 3 14 OSCILLATOR The internal oscillator generates the triangular waveform for the PWM charging and discharging with a constant current an internal capacitor. The switching frequency for each channel, FSW, is internally fixed at 150kHz so that the resulting switching frequency at the load side results in being tripled. The current delivered to the oscillator is typically 25A (corresponding to the free running frequency Fsw=150kHz) and it may be varied using an external resistor (ROSC) connected between the OSC pin and SGND or VCC (or a fixed voltage greater than 1.24V). Since the OSC pin is fixed at 1.24V, the frequency is varied proportionally to the current sunk (forced) from (into) the pin considering the internal gain of 6KHz/A. In particular connecting ROSC to SGND the frequency is increased (current is sunk from the pin), while connecting ROSC to VCC=12V the frequency is reduced (current is forced into the pin), according to the following relationships: kHz 1.237 7.422 10 R OSC vs. GND: F SW = 150kHz + ---------------------------- 6 ---------- = 150kHz + ---------------------------( k ) A R R ( k ) OSC OSC 6 12-1.237 - kHz 6.457 10 R OSC vs. 12V: F SW = 150kHz + ---------------------------- 6 ---------- = 150kHz + ---------------------------A R OSC ( k ) R OSC ( k ) 7 Maximum programmable switching frequency depends on the Current Reading Method selected. When reading across LS mosfet, the maximum switching frequency per phase must be limited to 500kHz to avoid current reading errors causing, as a consequence, current sharing errors. When reading across the inductor, higher switching frequency can be approached (device power dissipation must be checked prior to design high switching frequency systems). Figure 20. ROSC vs. Switching Frequency 14000 12000 10000 8000 6000 4000 2000 0 25 50 75 100 125 150 200 0 150 800 600 400 1000 250 350 450 550 650 750 850 ROSC [k] to 12V vs. Selected FSW [kHz] ROSC [k] to SGND vs. Selected FSW [kHz] 27/38 L6711 15 DRIVER SECTION The integrated high-current drivers allow using different types of power MOS (also multiple MOS to reduce the equivalent RdsON), maintaining fast switching transition. The drivers for the high-side mosfets use BOOTx pins for supply and PHASEx pins for return. The drivers for the low-side mosfets use VCCDRx pin for supply and PGNDx pin for return. A minimum voltage of 4.6V at VCCDRx pin is required to start operations of the device. VCCDRx pins must be connected together. The controller embodies a sophisticated anti-shoot-through system to minimize low side body diode conduction time maintaining good efficiency saving the use of Schottky diodes: when the high-side mosfet turns off, the voltage on its source begins to fall; when the voltage reaches 2V, the low-side mosfet gate drive is suddenly applied. When the low-side mosfet turns off, the voltage at LGATEx pin is sensed. When it drops below 1V, the high-side mosfet gate drive is suddenly applied. If the current flowing in the inductor is negative, the source of high-side mosfet will never drop. To allow the turning on of the low-side mosfet even in this case, a watchdog controller is enabled: if the source of the high-side mosfet doesn't drop for more than 240ns, the low side mosfet is switched on so allowing the negative current of the inductor to recirculate. This mechanism allows the system to regulate even if the current is negative. The BOOTx and VCCDRx pins are separated from IC's power supply (VCC pin) as well as signal ground (SGND pin) and power ground (PGNDx pin) in order to maximize the switching noise immunity. The separated supply for the different drivers gives high flexibility in mosfet choice, allowing the use of logic-level mosfet. Several combination of supply can be chosen to optimize performance and efficiency of the application. Power conversion input is also flexible; 5V, 12V bus or any bus that allows the conversion (See maximum duty cycle limitations) can be chosen freely. 16 POWER DISSIPATION Two main terms contribute in the device power dissipation: bias power and drivers' power. The first one depends on the static consumption of the device through the supply pins and it is simply quantifiable as follow: PDC = VCC * (ICC + 3 * ICCDRx + 3 * IBOOTx) Drivers' power is the power needed by the driver to continuously switch on and off the external mosfets; it is a function of the switching frequency and total gate charge of the selected mosfets. It can be quantified considering that the total power PSW dissipated to switch the mosfets (easy calculable) is dissipated by three main factors: external gate resistance (when present), intrinsic mosfet resistance and intrinsic driver resistance. This last term is the important one to be determined to calculate the device power dissipation. The total power dissipated to switch the mosfets results: PSW = 3 * (QG_HS * VBOOT + QG_LS * VCCDR) * FSW External gate resistors helps the device to dissipate the switching power since the same power PSW will be shared between the internal driver impedance and the external resistor resulting in a general cooling of the device.It is important to determine the device dissipated power in order to avoid the junction working beyond its maximum operative temperature. Moreover, since the device has an exposed pad to better dissipate the power, also the thermal resistance between junction and ambient is important. Figure 16 shows the Switching Power for different kind of mosfets driven. 28/38 L6711 Figure 21. Controller Power Dissipated (Quiescent + Switching; VCC = VCCDR = VBOOT = 12V) 3000 3000 Controller Dissiapted Power [mW] 2000 Controller Dissiapted Power [mW] 2500 2500 2000 1500 1500 QG_HS=36nC QG_LS=94nC 1000 QG_HS=72nC QG_LS=196nC QG_HS=48nC QG_LS=124nC QG_HS=24nC Q G_LS=62nC QG_HS=18nC Q G_LS=47nC 1000 QG_HS=48nC QG_LS=124nC 500 500 QG_HS=24nC QG_LS=62nC QG_HS=18nC QG_LS=47nC 700 900 0 100 QG_HS=72nC QG_LS=196nC 0 100 300 QG_HS=36nC QG_LS=94nC 500 300 500 700 900 Switching Frequency [kHz] per Phase Switching Frequency [kHz] per Phase RGATE = RGATE_MOSFET = 0 RGATE_HS = 2.2; RGATE_LS = 3.3; * RGATE_MOSFET = 1 17 SYSTEM CONTROL LOOP COMPENSATION The control loop is composed by the Current Sharing control loop and the Average Current Mode control loop. Each loop gives, with a proper gain, the correction to the PWM in order to minimize the error in its regulation: the Current Sharing control loop equalize the currents in the inductors while the Average Current Mode control loop fixes the output voltage equal to the reference programmed by VID. Figure 22 shows the block diagram of the system control loop. Figure 22. Main Control Loop Diagram L1 PWM1 1/5 CURRENT SHARING DUTY CYCLE CORRECTION L2 PWM2 1/5 1/5 L3 PWM3 IINFO1 IINFO2 IINFO3 ERROR AMPLIFIER VID IFB Co Ro 4/5 COMP FB ZF(s) RFB The average current mode control loop is reported in Figure 23. The current information IFB sourced by the FB pin flows into RFB implementing the dependence of the output voltage from the read current. The system can be modeled with an equivalent single phase converter which only difference is the equivalent inductor L/3 (where each phase has an L inductor).The ACM control loop gain results (obtained opening the loop after the COMP pin): 29/38 L6711 PWM Z F ( s ) ( RDROOP + Z P ( s ) ) G LOOP ( s ) = - -------------------------------------------------------------------------------------------------------------------ZF ( s ) 1 ( Z P ( s ) + Z L ( s ) ) -------------- + 1 + ----------- R FB A ( s ) A(s) Where: RSENSE is the mosfet RdsON or the Inductor DCR depending on the sensing element selected; R SENSE RDROOP = --------------------- RFB is the equivalent output resistance determined by the droop function; Rg ZP(s) is the impedance resulting by the parallel of the output capacitor (and its ESR) and the applied load Ro; ZF(s) is the compensation network impedance; ZL(s) is the parallel of the three inductor impedance; A(s) is the error amplifier gain; 4 V IN PWM = -- --------------- is the PWM transfer function where VOSC is the oscillator ramp amplitude and has 5 Vosc a typical value of 3V Removing the dependence from the Error Amplifier gain, so assuming this gain high enough, the control loop gain results: V IN ZF ( s ) Rs Z P ( s ) 4 G LOOP ( s ) = - -- ------------------ ------------------------------------ ------- + -------------- 5 V OSC Z P ( s ) + Z L ( s ) Rg RFB With further simplifications, it results: VIN Z ( s ) Ro + R DROOP 1 + s Co ( RDROOP //Ro + ESR ) 4 - FGLOOP ( s ) = - -- ------------------ -------------- ------------------------------------- --------------------------------------------------------------------------------------------------------------------------------5 V OSC RFB RL 2 L + s --------------- + Co ESR + Co RL + 1 L Ro + ----------s Co -3 3 3 3 Ro Considering now that in the application of interest it can be assumed that Ro>>RL; ESR< L6711 Figure 23. Equivalent Control Loop Gain Block Diagram (left) and Bode Diagram (right) ZF CF IFB RF RFB dB GLOOP VCOMP REF K ZF(s) PWM L/3 d*VIN Cout ESR VOUT LC Z T Rout 4 V 1 K = IN 5 VOSC RFB dB Compensation network can be simply designed placing Z=LC and imposing the cross-over frequency T as desired obtaining: L Co -RFB V OSC 5 L 3 RF = ---------------------------------- -- T ------------------------------------------------------CF = ------------------4 3 ( RDROOP + ESR ) V IN RF 18 LAYOUT GUIDELINES Since the device manages control functions and high-current drivers, layout is one of the most important things to consider when designing such high current applications. A good layout solution can generate a benefit in lowering power dissipation on the power paths, reducing radiation and a proper connection between signal and power ground can optimize the performance of the control loops. Integrated power drivers reduce components count and interconnections between control functions and drivers, reducing the board space. Here below are listed the main points to focus on when starting a new layout and rules are suggested for a correct implementation. 18.1 Power Connections. These are the connections where switching and continuous current flows from the input supply towards the load. The first priority when placing components has to be reserved to this power section, minimizing the length of each connection and loop as much as possible. To minimize noise and voltage spikes (EMI and losses) these interconnections must be a part of a power plane and anyway realized by wide and thick copper traces: loop must be anyway minimized. The critical components, i.e. the power transistors, must be located as close as possible one to the other. Figure 24 shows the details of the power connections involved and the current loops. The input capacitance (CIN), or at least a portion of the total capacitance needed, has to be placed close to the power section in order to eliminate the stray inductance generated by the copper traces. Low ESR and ESL 31/38 L6711 capacitors are preferred. Use as much VIAs as possible when power traces have to move between different planes on the PCB: this reduces both parasitic resistance and inductance. Moreover, reproducing the same high-current trace on more than one PCB layer will reduce the parasitic resistance associated to that connection. Connect output bulk capacitor as near as possible to the load, minimizing parasitic inductance and resistance associated to the copper trace also adding extra decoupling capacitors along the way to the load when this results in being far from the bulk capacitor bank. Figure 24. Power connections and related connections layout guidelines (same for all phases). To limit CBOOT Extra-Charge VIN BOOTx VIN CBOOT UGATEx PHASEx CIN L CIN L PHASEx VCC LGATEx PGNDx LOAD SGND +Vcc LOAD a. PCB power and ground planes areas b. PCB small signal components placement 18.2 Power Connections Related. Figure 24 shows some small signal components placement. Gate and phase traces must be sized according to the driver RMS current delivered to the power mosfet. The device robustness allows managing applications with the power section far from the controller without losing performances. Anyway, when possible, it is suggested to minimize the distance between controller and power section. In addition, since the PHASEx pin is the return path for the high side driver, this pin might be connected directly to the High Side mosfet Source pin to have a proper driving for this mosfet. For the LS mosfets, the return path is the PGNDx pin: it can be connected directly to the power ground plane. Bootstrap capacitor must be placed as close as possible to the BOOTx and PHASEx pins to minimize the loop that is created. Decoupling capacitor from VCC and SGND placed as close as possible to the involved pins. Decoupling capacitor from VCCDRx and PGNDx placed as close as possible to those pins. This capacitor sustains the peak currents requested by the low-side mosfet drivers. Sensible components must be referred to SGND (when present): frequency set-up resistor ROSC, offset resistor ROFFSET, TC resistor RTC and OVP resistor ROVP. Star grounding: Connect SGND to PGND plane in a single point to avoid that drops due to the high current delivered causes errors in the device behavior. An additional ceramic capacitor is suggested to place near HS mosfet drain. This helps in reducing HF noise. VSEN pin filtered vs. SGND helps in reducing noise injection into device. OUTEN pin filtered vs. SGND helps in reducing false trip due to coupled noise: take care in routing driving net for this pin in order to minimize coupled noise. PHASE pin spikes. Since the HS mosfet switches hardly, heavy voltage spikes can be observed on the PHASEx pins. If these voltage spikes overcome the max breakdown voltage of the pin, the device can 32/38 L6711 absorb energy and it can cause damages. The voltage spikes must be limited by proper layout; by the use of gate resistors, Schottky diodes in parallel to the low side mosfets and/or snubber network on the low side mosfets, and cannot overcome 26V, for 20nSec, at FSW = 600kHz. Boot Capacitor Extra Charge. Systems that do not use Schottky diodes might show big negative spikes on the phase pin. This spike can be limited as well as the positive spike but has an additional consequence: it causes the bootstrap capacitor to be over-charged. This extra-charge can cause, in the worst case condition of maximum input voltage and during particular transients, that boot-to-phase voltage overcomes the abs. max. ratings also causing device failures. It is then suggested in this cases to limit this extra-charge by adding a small resistor in series to the boot diode (one resistor can be enough for all the three diodes if placed upstream the diode anode, see Figure 24). 18.3 Current Sense Connections. Remote Buffer: The input connections for this component must be routed as parallel nets from the FBG/ FBR pins to the load in order to compensate losses along the output power traces and also to avoid the pick-up of any common mode noise. Connecting these pins in points far from the load will cause a nonoptimum load regulation, increasing output tolerance. Current Reading: The Rg resistors have to be placed as close as possible to the CSx- and CSx+ pins in order to limit the noise injection into the device; this is still valid also for the Rg(RC)-Cg network used when sensing current across the inductor. The PCB traces connecting these resistors to the reading point must use dedicated nets, routed as parallel traces in order to avoid the pick-up of any common mode noise. Figure 25. Device orientation (left) and sense nets routing (right: red for Lsense, black for LSsense) HS1 HS2 HS3 LS1 LS2 LS3 R C R R R R R R C Vias to PGND plane VCCDR It's also important to avoid any offset in the measurement and, to get a better precision, to connect the traces as close as possible to the sensing elements. Symmetrical layout is also suggested. Small filtering capacitor can be needed between VOUT and SGND on the CSx- line, placed near the controller, allowing higher layout flexibility in the current sense connection. Vias to PGND plane CS3+ CS2+ CS1+ CS3- CS2- CS1- (or Sense Resistor) (or Sense Resistor) To LS Mosfet To regulated Output To Inductor 33/38 L6711 19 EMBEDDING L6711-BASED VRDS... When embedding the VRD into the application, additional care must be taken since the whole VRD is a switching DC/DC regulator and the most common system in which it has to work is a digital system such as MB or similar. In fact, latest MB has become faster and powerful: high speed data bus are more and more common and switching-induced noise produced by the VRD can affect data integrity if not following additional layout guidelines. Few easy points must be considered mainly when routing traces in which switching high currents flow (switching high currents cause voltage spikes across the stray inductance of the traces causing noise that can affect the near traces): Keep safe guarding distance between high current switching VRD traces and data buses, especially if high-speed data bus to minimize noise coupling. Keep safe guard distance or filter properly when routing bias traces for I/O sub-systems that must walk near the VRD. Possible causes of noise can be located in the PHASE connections, Mosfet gate drive and Input voltage path (from input bulk capacitors and HS drain). Also PGND connections must be considered if not insisting on a power ground plane. These connections must be carefully kept far away from noise-sensitive data bus. Since the generated noise is mainly due to the switching activity of the VRM, noise emissions depend on how fast the current switch. To reduce noise emission levels, it is also possible, in addition to the previous guidelines, to reduce the current slope and then to increase the switching times: this will cause, as a consequence of the higher switching time, an increase in switching losses that must be considered in the thermal design of the system. 34/38 L6711 20 TQFP48 Mechanical Data & Package Dimensions Figure 26. TQFP48 Mechanical Data & Package Dimensions mm DIM. MIN. TYP. MAX. MIN. TYP. MAX. inch A A1 A2 b c D D1 D2 D3 e E E1 E2 E3 e L L1 k ccc 0.45 8.80 6.80 2.00 5.50 0.50 0.60 1.00 0.05 0.95 0.17 0.09 8.80 6.80 2.00 5.50 0.50 9.00 7.00 9.00 7.00 1.00 0.22 1.20 0.15 1.05 0.27 0.20 9.20 7.20 4.25 0.002 0.037 0.006 0.004 0.346 0.268 0.079 0.217 0.020 9.20 7.20 4.25 0.346 0.268 0.079 0.217 0.019 0.75 0.018 0.024 0.039 0.354 0.276 0.354 0.276 0.039 0.008 0.047 0.006 0.041 0.010 0.008 0.362 0.283 0.167 OUTLINE AND MECHANICAL DATA 0.362 0.283 0.167 Body: 7 x 7 x 1.0mm 0.030 0(min.), 3.5(typ.), 7(max.) 0.08 0.0031 TQFP48 - EXPOSED PAD 7222746 B 35/38 L6711 Table of Contents 1 2 3 4 5 FEATURES ........................................................................................................................................1 APPLICATIONS .................................................................................................................................1 DESCRIPTION...................................................................................................................................1 DEVICE DESCRIPTION ..................................................................................................................13 CURRENT READING AND CURRENT SHARING CONTROL LOOP ............................................13 5.1 LOW SIDE CURRENT READING..........................................................................................14 5.2 INDUCTOR CURRENT READING ........................................................................................15 DAC SELECTION ............................................................................................................................16 REMOTE VOLTAGE SENSE...........................................................................................................16 7.1 WARNING:.............................................................................................................................16 VOLTAGE POSITIONING................................................................................................................17 8.1 DROOP FUNCTION ..............................................................................................................17 8.2 OFFSET .................................................................................................................................18 8.3 INTEGRATED THERMAL SENSOR......................................................................................19 DYNAMIC VID TRANSITION ...........................................................................................................20 9.1 VID_SEL = OPEN (SEE FIGURE 14). ...................................................................................21 9.1.1 WARNING: .....................................................................................................................21 9.2 VID_SEL = GND (SEE FIGURE 15). .....................................................................................22 ENABLE AND DISABLE ..................................................................................................................22 10.1 OUTEN PIN............................................................................................................................22 10.2 NOCPU (VID [0;5]=11111X) ..................................................................................................22 SOFT START ...................................................................................................................................23 OUTPUT VOLTAGE MONITOR AND PROTECTIONS ...................................................................23 12.1 UVP PROTECTION ...............................................................................................................23 12.2 PROGRAMMABLE OVP PROTECTION ...............................................................................23 12.3 PRELIMINARY OVP PROTECTION (PRE-OVP) ..................................................................23 OVERCURRENT PROTECTION .....................................................................................................24 13.1 LS MOSFET SENSE (CS_SEL=OPEN) OVERCURRENT ...................................................25 13.2 TON LIMITED OUTPUT VOLTAGE.......................................................................................25 13.2.1 CONSTANT CURRENT OPERATION ...........................................................................25 13.3 INDUCTOR SENSE (CS_SEL = SGND) OVER CURRENT..................................................26 OSCILLATOR...................................................................................................................................27 DRIVER SECTION...........................................................................................................................28 POWER DISSIPATION ....................................................................................................................28 SYSTEM CONTROL LOOP COMPENSATION...............................................................................29 LAYOUT GUIDELINES ....................................................................................................................31 18.1 POWER CONNECTIONS. .....................................................................................................31 18.2 POWER CONNECTIONS RELATED. ...................................................................................32 18.3 CURRENT SENSE CONNECTIONS.....................................................................................33 EMBEDDING L6711-BASED VRDS................................................................................................34 TQFP48 MECHANICAL DATA & PACKAGE DIMENSIONS ...........................................................35 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 36/38 L6711 Table 8. Revision History Date June 2004 November 2004 Revision 1 2 First Issue Description of Changes Modificated the Paragraph 18.2 on the page 32/38. 37/38 L6711 Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a registered trademark of STMicroelectronics. All other names are the property of their respective owners (c) 2004 STMicroelectronics - All rights reserved STMicroelectronics GROUP OF COMPANIES Australia - Belgium - Brazil - Canada - China - Czech Republic - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan Malaysia - Malta - Morocco - Singapore - Spain - Sweden - Switzerland - United Kingdom - United States www.st.com 38/38 |
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